Chuck,
If you are unsure of the output impedance, one solution is to add a low noise buffer (op amp), with a 50 ohm series output resistor and large DC blocking capacitor (passing 600 Hz would require about 22 uF to minimize impact on amplitude accuracy). You could also add some gain here depending on the expected signal level.

If this is not an option, you could try measuring the output impedance, or alternately see how much the amplitude changes when you add a e.g. 100 ohm series resistor to your output, and then use standard resistor divider equations to solve for output impedance (assuming a 50 ohm input impedance).

The goal of VBW > RBW is essentially to remove any significant filtering after the RBW. Our software is a little different in that we bypass VBW processing unless VBW < RBW or the average detector is used. So for Spike software, VBW equal to RBW is what you want.

I assume you are doing FCC 15.247 “The maximum permitted peak conducted output power is +30 dBm (1 W). However, the power spectral density conducted from the intentional radiator to the antenna shall not be greater than +8 dBm in any 3 kHz band during any time interval of continuous transmission. ”

To achieve this, I would use the following settings:
1) Spur reject off
2) Video detector set to min/max
3) 3 kHz CISPR / Gaussian RBW, 3 kHz VBW (in Spike, when VBW = RBW and min/max selected, there is no video filtering)
4) Trace set to max hold
After accumulating for several seconds, peak search. This should give you the correct reading. Let me know if I misunderstood the test you are running.

If you need triggered zero span (I/Q) acquisitions, I highly recommend the BB60C. Triggering on the SA44B is very limited and it is easy to lock up the device if you are not very careful.

We intentionally do not trigger sweeps, to avoid the situation where people try to sync a swept signal generator with our spectrum analyzer, because a stepped FFT-based spectrum analyzer, even perfectly synced, is only actively looking at the spectrum a fraction of the time.

The solution is to use our tracking generator, which has firmware to switch frequency at appropriate times for the architecture.

Bruce,
If you have not rebooted your computer since installing the driver, try that first. If you have, we can try the CDM uninstaller, to remove any history of the SA44B and get a fresh start.

http://www.ftdichip.com/Support/Utilities/CDMUninstaller_v1.4.zip

Unplug the SA44B, and then run the CDMUninstallerGUI
Use VID = 0403, PID = 6010, then click “Add” then “Remove Devices”
Plug in your SA44B and wait 3 minutes. Watch the LED. It should turn green once the driver finishes installing.
Let me know if this helps.

I don’t know if this will help, but making sure the driver is up to date may help. FTDI puts out a nice guide with some helpful troubleshooting tips: http://www.ftdichip.com/Support/Documents/InstallGuides/AN_396%20FTDI%20Drivers%20Installation%20Guide%20for%20Windows%2010.pdf

Raash,
Using just the open (or short) is suboptimal, and the directionality of your coupler limits the accuracy of your VSWR measurement. But it’s my understanding that correcting with open / short / load would require phase information, which we cannot acquire with a scalar system.

Raash,
For the Smith chart utility, the time bases must be locked, but the sync signal is not used. Only one BNC is required.
Connect the TG 10 MHz out to the SA external reference in, then enable external reference in the software. Note that on the SA series, with external reference enabled, do not remove the reference signal until after you close the software.

That sounds like an ambitious project. Off the top of my head, it sounds doable. You’d have to generate a CSV file with your two tones, measure the amplitude and phase of the 3rd order responses (in a highly linear measurement system), and then subtract the 3rd order responses from your CSV file (or something similar). I would expect that you could drop your 3rd order intermod products by 20 dB at least, but you’d have to try it to know. And it would vary with temperature and frequency, so you would probably want to do it right before using it or after a 30 minute warmup. Assuming you want tones spaced 2 MHz (+/- 1 MHz) you’d want to choose a DAC clock that is a multiple of 6 * tone spacing. Something like 96 MHz would probably work nicely.
If you can’t get phase info, you could use the amplitude of the IM products and hunt for the right phase that exactly cancels the undesired IM products.

I see you’ve put some thought into this.
1) The RBW calculation you gave is actually the bin size calculation. RBW is bin size * FFT window bandwidth. Typically, a window bandwidth of about 2 is required or spectrum analysis, for a 30k RBW.
2) The bin size will be as the power-of-2 math determines. Rarely a nice number.
3) We will have to drop to 16 bit amplitude data for these sweeps, typically 0.01 dB resolution.
4) The 120 us includes tune time of about 20 us.

Sure. The SM200A has the ability to perform FFTs on about 500 million data points per second, but the maximum FFT size in our design is 16384. The dual channel 500 MSPS ADC is run through a FIR correction filter, a digital single sideband select filter, and then converted to 250 MSPS I/Q data (160 MHz). This can then be processed with the on-board FFT engine, or decimated further to 50 MSPS (40 MHz) and streamed to the PC, where much larger FFTs are possible.
Yes, the limitation for narrower RBWs is the USB3 speed. For FFTs on the FPGA, we impose a 120 microsecond minimum dwell time at each LO, which becomes the limiting factor, but for a 30k RBW, USB3 speed is nearly a limiting factor even for basic amplitude data.

You also mentioned sweep speed at 1 kHz RBW. While we should have 1 THz/sec at 30 kHz RBW, once you cross into the 40 MHz patches, this slows down. You could expect about a 2-4 ms acquisition time per patch for a 1 kHz RBW. Assuming your CPU was decent, this would give you 40 MHz / 4 ms, or about 10 GHz/sec (maybe ~2 seconds for a 20 GHz sweep)

Mikes,
We have been using the 2/3 (IP3-DANL@1Hz RBW) for dynamic range.
You mentioned a -130 dBm signal with a -20 dBm signal. The noise floor for a -20 dBm signal and reference level would be about -157 dBm/Hz, so I’d say you’d have to have a 100 Hz RBW to clearly see the -130 dBm signal, and you’d have to ensure phase noise wasn’t a limiting factor. At 1 GHz, internal phase noise would not block a -110 dBc signal if you were > 300 Hz from a clean -20 dBm CW blocker. The phase noise should be better than -133 dBc/Hz (>= 10k offset) at 1 GHz, but this goes down 20 dB / decade as you increase frequency.
Also, for 160 MHz patches, you are limited to >= 30 kHz RBW. For lower RBW, 40 MHz patches are used.

David,
It sounds like whenever it tries to download the calibration table, it hangs. We may need to take a look at it here. You can contact support at signalhound dot com for an RMA. I apologize for the inconvenience. If you purchased it less than 2 years ago, it should be covered under warranty.

Thank you for the feedback and tips!

The SA44B / TG44A with coupler can be used to measure magnitude S11 and magnitude S21. It cannot provide the phase of S11 or S21. This is the critical difference between a VNA and SNA.

A two port VNA can also provide S22 and S12 without swapping your device around.

The best way to mute the output is to set the frequency to 0 Hz, amplitude to -30 dBm.

Eduk,
We have observed, when the preamp is used, at 0 dB of attenuation, it can take a long time for the DC blocking cap to reach steady state (up to 30 seconds). Typically, setting your reference level to -30 dBm briefly (preamp on, attenuator 5-10 dB), before proceeding to maximum sensitivity, allows quick settling.

On a fraction of units, the preamp’s DC blocking cap never reaches steady state without a better DC path to ground. We typically have to add a 2.2k resistor to help bleed this off. You can contact us for a warranty repair if your SA44B is exhibiting this behavior. I apologize for the inconvenience.

We will be updating the design to add this resistor in the near future.

The SA44B definitely has some shortcomings when it comes to EMC testing. It relies on a software image rejection algorithm to get rid of both image responses and its own spurious responses. This algorithm tends to work better on sparse spectrum with narrow RBWs, where the chance of a spur and an image overlapping is low. When there are many signals closely spaced, especially at low frequency where there may be higher frequencies present mixing with harmonics of the LO, the chance of spurious signals overlapping on the primary and image sweep are high.

This is one of the reasons we strongly recommend the BB60C for EMC work. With hardware image rejection and specified spurious performance, you get reliable EMC sweeps.

We aren’t going to IWCE, but we will be at IMS in June.

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